Distortion reduction calibration

ABSTRACT

Techniques are disclosed for compensating for second-order distortion in a wireless communication device. In a zero-intermediate frequency (IF) or low-IF architecture, IM2 distortion generated by the mixer ( 20 ) results in undesirable distortion levels in the baseband output signal. A compensation circuit ( 104 ) provides a measure of the IM2 distortion current independent of the radio frequency (RF) pathway to generate an IM2 calibration current. The IM2 calibration current is combined with the baseband output signal to thereby eliminate the IM2 currents generated within the RF pathway. In one embodiment, the calibration is provided at the factory during final testing. In alternative embodiment, additional circuitry ( 156, 158 ) may be added to the wireless communication device to provide a pathway between the transmitter ( 150 ) and the receiver ( 146 ). The transmitter signal is provided to the receiver to permit automatic calibration of the unit. An internal signal source ( 162 ) may be used in place of the transmitter ( 150 ). The auto-calibration may be performed to eliminate IM2 distortion or permit optimization of the circuit to minimize other forms of distortion or interference.

CROSS REFERENCE TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.10/066,115, filed Feb. 1, 2002, now U.S. Pat. No. 7,657,241 entitled“DISTORTION REDUCTION CALIBRATION”.

BACKGROUND

1. Field of the Invention

The present invention is related generally to wireless communicationdevices, and, more particularly, to a system and method for a distortionreduction calibration circuit in a wireless communication device.

2. Description of the Related Art

Wireless communication systems are proliferating as more and moreservice providers add additional features and technical capabilities. Alarge number of service providers now occupy a relatively limitedportion of the radio frequency spectrum. Due to this crowding, increasedinterference between wireless communication systems is commonplace. Forexample, wireless communication systems from two different serviceproviders may occupy adjacent portions of the spectrum. In thissituation, interference may be likely.

One example of such interference occurs in a code division multipleaccess (CDMA) wireless system. In one embodiment, a CDMA system occupiesa portion of the frequency spectrum adjacent to a portion of thefrequency spectrum allocated to a conventional cellular telephonesystem, sometimes referred to as an advanced mobile phone system (AMPS).

Conventional CDMA units attempt to eliminate undesirable signals byadding filters following the mixer stage. FIG. 1 illustrates one knownimplementation of a direct-to-baseband or low IF wireless system 10 inwhich a radio frequency (RF) stage 12 is coupled to an antenna 14. Theoutput of the RF stage 12 is coupled to an amplifier 16, which amplifiesthe radio frequency signals. It should be noted that the RF stage 12 andthe amplifier 16 may include conventional components such as amplifiers,filters, and the like. The operation of these stages is well known andneed not be described in greater detail herein.

The output of the amplifier 16 is coupled to a splitter 18 that splitsthe processed signal into two identical signals for additionalprocessing by a mixer 20. The splitter 18 may be an electronic circuitor, in its simplest form, just a wire connection. The mixer 20 comprisesfirst and second mixer cores 22 and 24, respectively. The mixers 22 and24 are identical in nature, but receive different local oscillatorsignals. The mixer core 22 receives a local oscillator signal,designated LOI, while the mixer core 24 receives a local oscillatorsignal, designated as LOQ. The local oscillator signals are 90° out ofphase with respect to each other, thus forming a quadrature mixer core.The output of the mixer 20 is coupled to jammer rejection filter stage26. Specifically, the output of the mixer core 22 is coupled to a jammerrejection filter 28 while the output of the mixer core 24 is coupled toa jammer rejection filter 30. The operation of the jammer rejectionfilters 28 and 30 is identical except for the quadrature phaserelationship of signals from the mixer 20. The output of the jammerrejection filters 28 and 30 are the quadrature output signals I_(OUT)and Q_(OUT) respectively.

The jammer rejection filters 28 and 38 are designed to remove unwantedsignals, such as signals from transmitters operating at frequencies nearthe frequency of operation of the system 10. Thus, the jammer rejectionfilters 28 and 30 are designed to remove “out-of-band” signals. Inoperation, the jammer rejection filters 28 and 30 may be lowpassfilters, bandpass filters, or complex filters (e.g., a single filterwith two inputs and two outputs), depending on the implementation of thesystem 10. The operation of the jammer rejection filters 28 and 30 arewell known in the art and need not be described in greater detailherein. While the jammer rejection filters 28 and 30 may minimize theeffects of out-of-band signals, there are other forms of interferencefor which the jammer rejection filters are ineffective.

For example, distortion products created by the mixer 20 may result ininterference that may not be removed by the jammer rejection filters 28and 30. If one considers a single CDMA wireless unit, that unit isassigned a specific radio frequency or channel in the frequencyspectrum. If an AMPS system is operating on multiple channels spacedapart from each other by a frequency Δω_(J), then the second-orderdistortion from the mixer 20 will create a component at a frequencyΔω_(J) in the output signal. It should be noted that the second orderdistortion from the mixer 20 will create signal components at the sumand difference of the two jammer frequencies. However, the signalresulting from the sum of the jammer frequencies is well beyond theoperational frequency of the wireless device and thus does not causeinterference. However, the difference signal, designated herein asΔω_(J), may well be inside the desired channel and thus causesignificant interference with the desired signal.

In this circumstance, the AMPS signals are considered a jammer signalssince they create interference and therefore jam the desirable CDMAsignal. Although the present example refers to AMPS signals as jammersignals, those skilled in the art will appreciate that any other radiofrequency sources spaced at a frequency of Δω_(J) from each other may beconsidered a jammer.

If this second-order distortion signal is inside the channel bandwidth,the jammer rejection filters 28 and 30 will be ineffective and theresultant interference may cause an unacceptable loss ofcarrier-to-noise ratio. It should be noted that this interference mayoccur regardless of the absolute frequencies of the jammer signals. Onlythe frequency separation is important if the second-order distortionresults in the introduction of an undesirable signal into the channelbandwidth of the CDMA unit.

Industry standards exist that specify the level of higher orderdistortion that is permitted in wireless communication systems. A commonmeasurement technique used to measure linearity is referred to as aninput-referenced intercept point (IIP). The second order distortion,referred to as IIP2, indicates the intercept point at which the outputpower in the second order signal intercepts the first order signal. Asis known in the art, the first order or primary response may be plottedon a graph as the power out (P_(OUT)) versus power in (P_(IN)). In alinear system, the first order response is linear. That is, the firstorder power response has a 1:1 slope in a log-log plot. The power of asecond order distortion product follows a 2:1 slope on a log-log plot.It follows that the extrapolation of the second order curve willintersect the extrapolation of the fundamental or linear plot. Thatpoint of intercept is referred to as the IIP2. It is desirable that theIIP2 number be as large as possible. Specifications and industrystandards for IIP2 values may vary from one wireless communicationsystem to another and may change over time. The specific value for IIP2need not be discussed herein.

It should be noted that the second-order distortion discussed herein isa more serious problem using the direct down-conversion architectureillustrated in FIG. 1. In a conventional super-heterodyne receiver, theRF stage 12 is coupled to an intermediate frequency (IF) stage (notshown). The IF stage includes bandpass filters that readily remove thelow frequency distortion products. Thus, second-order distortion is nota serious problem with a super-heterodyne receiver. Therefore, the IIP2specification for a super-heterodyne receiver is generally not difficultto achieve. However, with the direct down-conversion receiver, such asillustrated in FIG. 1, any filtering must be done at the basebandfrequency. Since the second-order distortion products at the frequencyseparation, Δω_(J), regardless of the absolute frequency of the jammers,the IIP2 requirements are typically very high for a direct-conversionreceiver architecture. The IIP2 requirement is often the single mostdifficult parameter to achieve in a direct down-conversion receiverarchitecture.

As noted above, the second-order distortion is often a result ofnon-linearities in the mixer 20. There are a number of factors that leadto imbalances in the mixer 20, such as device mismatches (e.g.,mismatches in the mixer cores 22 and 24), impedance of the localoscillators, and impedance mismatch. In addition, factors such as theduty cycle of the local oscillator also has a strong influence on thesecond-order distortion. Thus, the individual circuit components andunique combination of circuit components selected for a particularwireless communication device results in unpredictability in the IIP2value for any given unit. Thus, calibration of individual units may berequired to achieve the IIP2 specification.

Therefore, it can be appreciated that there is a significant need for asystem and method for wireless communication that reduces theundesirable distortion products to an acceptable level. The presentinvention provides this and other advantages as will be apparent fromthe following detailed description and accompanying figures.

SUMMARY

Novel techniques are disclosed for distortion reduction calibration. Inan exemplary embodiment, a distortion reduction circuit for use in awireless communication device has a radio frequency (RF) receiver andcomprises a gain stage having an input coupled to the receiver and anoutput with the gain stage controlling an amplitude of an output signalrelated to a second order nonlinear response within the receiver. Anoutput coupling circuit couples the gains stage output to the receiver.

In one embodiment, the gain stage amplitude control is based on theamplitude of the second order nonlinear response within the receiver.The signal related to the second order nonlinear response within thereceiver may be inherently generated by circuitry within the receiver ormay be generated by a squaring circuit coupled to the receiver.

When implemented with an RF receiver generating a down-converted outputsignal, the output coupling circuit may comprise an adder having firstand second inputs with the first input configured to receive the outputsignal from the receiver and the second input configured to receive thegain stage output signal. The gain stage may generate an output currentrelated to the second order nonlinear response within the receiver. Theoutput coupling circuit may be a direct connection to the down-convertedoutput signal of the receiver.

In one embodiment, the circuit is for use in a factor calibrationwherein the receiver generates a down-converted output signal and isconfigured to receive an external input signal to permit the adjustmentof the gain stage to thereby minimize the second order nonlinearresponse of the receiver output signal.

In another embodiment, an automatic calibration circuit may be used withthe wireless communication device wherein a signal source generates atest signal and a switch is selectively activated to couple the signalsource to a receiver input terminal to couple the test signal to thereceiver input terminal and thereby permit distortion reductionadjustments on the receiver.

The switch circuit maybe selectively activated in an auto-calibrationmode or activated at predetermined times.

In one embodiment, the signal source comprises an internal signalgenerator. The internal signal generator may generate the test signalhaving multiple frequency components having a predetermined spectralspacing. In another embodiment, the wireless communication deviceincludes an RF transmitter and the circuit may further comprise atransmitter control to control an input signal to the transmitter andselectively activated during the auto-calibration process to generatethe test signal. In one embodiment, the circuit may further include anattenuator coupled to a transmitter output terminal to generate anattenuated output signal as the test signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a functional block diagram of a conventional wirelesscommunication receiver.

FIG. 2 is a functional block diagram of a generic implementation of thepresent invention.

FIG. 3 is a functional block diagram of a receiver mixer illustratingone implementation of the present invention.

FIG. 4 is a schematic diagram illustrating one possible implementationof the present invention.

FIG. 5 is a functional block diagram of an alternative implementation ofthe present invention.

FIG. 6 is a functional block diagram of another alternativeimplementation of the present invention.

DETAILED DESCRIPTION

The present invention is directed to a calibration circuit and methodthat simplifies the calibration process for individual wirelesscommunication devices. The term “wireless communication device”includes, but is not limited to, cellular telephones, personalcommunication system (PCS) devices, radio telephones, mobile units, basestations, satellite receivers and the like. In one embodiment, thecalibration circuit is used at assembly to compensate for variations incomponents. In an alternative embodiment, also described herein, anonboard calibration circuit can be used to compensate for componentmismatch due to circuit aging or other changes in circuit operationalparameters.

IIP2 performance presents a major challenge in direct conversiondown-converters. The required values of IIP2 are usually very high andthe actual performance tends to be difficult to predict because it isalmost exclusively determined by statistical phenomena. That is,component mismatch tends to be a statistical phenomena. Even so-called“matched” components on an integrated circuit are subject to variationsin operating characteristics due to processing variations of anintegrated circuit. Similarly, external components are also subject tovariation that is unpredictable and cannot be readily accounted for indesigning a radio frequency (RF) circuit.

There are some known techniques for suppressing IIP2 distortion, butthese processes tend to be complicated or introduce new spurs (i.e.,undesirable frequency components) and require a change in frequency plan(i.e., reallocation of the frequency spectrum). In addition, these knowntechniques interfere with the RF path and will degrade other RFparameters such as noise figure and IIP3. As a result, these knowncircuits lead to more complicated circuitry, increased cost, anddecreased performance.

In contrast, the present invention uses a feed-forward technique, whichrelies on a one-time calibration at the phone level. The circuitry ofthe present invention is designed such that it does not interfere withthe RF path, and the RF path can therefore be optimized for other RFperformance parameters (e.g., noise figure and IIP3), independently ofIIP2. All of the calibration works at baseband frequencies, whichfacilitates the design and layout and enables lower power consumption.

As previously discussed, the second order nonlinear distortion is asignificant problem in direct conversion receiver architectures (i.e.,zero IF or low IF architectures). While heterodyne receiverarchitectures also generate second order distortion, other conventionaltechniques may be used to reduce the unwanted nonlinear distortion. Forexample, careful selection of the IF frequency followed by IF filteringmay typically reduce the second order nonlinear distortion to anacceptable level in heterodyne receivers. While the discussion hereinuses low IF or zero IF examples, the principles of the present inventionmay be applied to other receiver architectures, including heterodynereceivers.

Furthermore, the description presented herein may refer to a basebandsignal, resulting from a low IF or zero IF mixing. However theprinciples of the present invention apply generally to a down-convertedsignal that is generated by a mixer. Therefore, the present invention isnot limited by the receiver architecture, but can generally be appliedto any down-converted signal having a second order nonlinear distortion.

The present invention is embodied in a system 100, which is shown in anexemplary form in the functional block diagram of FIG. 2. The system 100processes an RF_(in) signal, which is illustrated in FIG. 2 in the formof a voltage (V_(RF)). The RF_(in) signal is processed by a conventionalRF block 102. The RF block may include amplifiers, filters, and thelike. In addition, the RF block typically includes a mixer, such as themixer 20 illustrated in FIG. 1 to convert the RF signal to a basebandsignal. As illustrated in FIG. 2, the baseband signal comprisescomponents that are identified as i_(BBdesired)+i_(IM2). This isintended to represent the desired baseband signal combined with theundesirable signal resulting from second order distortion within the RFblock 102.

The system 100 also includes a compensation branch 104, which comprisesa squaring circuit 106, lowpass filter 108, and variable gain amplifier(VGA) 110. The squaring circuit 106 provides a squared version of thevoltage V_(RF). As those skilled in the art will appreciate, thesquaring circuit produces a number of undesirable harmonics at multiplefrequencies. The low pass filter 108 is designed to eliminate theundesirable frequencies so that the compensation branch 104 does notproduce undesirable interference. The VGA 110 is used to attenuate oramplify a compensation signal identified in FIG. 2 as i_(IM2cal). Thecompensation signal i_(IM2cal) is combined with the output of the RFblock by an adder 114. The output of the adder 114 is the desired signali_(outBB). If the compensation current i_(IM2cal) equals the undesirablesignal component i_(IM2), the output signal i_(outBB) equals the desiredsignal i_(BBdesired).

As illustrated in FIG. 2, the IM2 calibration scheme relies on cancelingthe IM2 output current generated by the RF block 102 with a programmableIM2 current derived from another source. In the present example, theprogrammable compensation current is derived directly from the RFsignal, but does not interact with the RF pathway in the RF block 102.Thus, the advantage of this technique is that it does not interfere withthe RF path. Therefore, the introduction of IM2 calibration will notdegrade other RF parameters such as gain, noise figure and IIP3.

For proper cancellation of the undesirable signal by the adder 114, thetwo IM2 currents (i.e., i_(IM2) and i_(IM2cal)) must either be in-phaseor 180 degrees out of phase. Due to the mechanism generating IM2, thisis expected to be the case and will be derived below. As noted above,the RF block 102 contains conventional components, such as the mixer 20(see FIG. 1). The IM2 current generated by the mixer 20 can be expressedin the form:i _(IM2mix)(t)=α_(2mix) ·V _(RF)(t)²  (1)

Expressing V_(RF) in polar form and taking into account that it may beattenuated by some factor α_(mix) and phase-shifted by some phaseφ_(mix) through the mixer circuitry, we obtain:i _(IM2mix)(t)=α_(2mix)·(α_(mix) ·A(t)cos(ω_(RF) ·t+φ(t)+φ_(mix)))²  (2)and expanding this yields

$\begin{matrix}{{i_{IM2mix}(t)} = {{\frac{1}{2} \cdot a_{2{mix}} \cdot a_{mix}^{2} \cdot \mspace{130mu}{A(t)}^{2}}\left( {1 + {\cos\left( {{2{\omega_{RF} \cdot t}} + {2\;{\phi(t)}} + {2\phi_{mix}}} \right)}} \right)}} & (3)\end{matrix}$

A portion of the signal represented by equation (3) is dependent on avalue 2ω_(RF). This portion of equation (3) is of little concern in thisanalysis since it is very high frequency and will be filtered away usingconventional techniques. However, the low-frequency part could landinside the desired baseband channel. So the IM2 product of interest fromequation (3) is

$\begin{matrix}{{i_{IM2mixLF}(t)} = {\frac{1}{2} \cdot a_{2{mix}} \cdot a_{mix}^{2} \cdot {A(t)}^{2}}} & (4)\end{matrix}$

Similarly, the IM2 compensation current generated at the output of theVGA 110 in FIG. 2 is given by

$\begin{matrix}{{i_{IM2cal}(t)} = {\frac{1}{2} \cdot a_{cal} \cdot {A(t)}^{2}}} & (5)\end{matrix}$

where α_(cal) is a programmable scaling factor. Cancellation of IM2 bythe adder 114 is achieved whenα_(cal)=−α_(2mix)·α_(mix) ²  (6)

Thus, IM2 cancellation should be possible independently of the RF phaseshift φ_(mix) through the mixer.

In a typical implementation of the RF block 102, the mixer cores are themain IM2 contributors. Therefore, to improve tracking between the IM2source (i.e., the mixer core) and the IM2 calibration signal, it wouldbe desirable to derive the IM2 calibration signal from the mixer coresthemselves. This is fortunately straight-forward, because theemitter-nodes of the mixer core present a strong second-ordernon-linearity. Conceptually, the IM2 calibration circuit can beimplemented as shown in the functional block diagram of FIG. 3. For thesake of clarity, FIG. 3 illustrates only a single mixer core (i.e.,either the I mixer or the Q mixer core). Those skilled in the art willrecognize that an additional mixer core and calibration circuit areimplemented in accordance with the description provided herein. Itshould also be noted that the simplified functional block diagram ofFIG. 2 represents a single ended system while the functional blockdiagram of FIG. 3 is a differential implementation with differentialinputs and differential outputs. Those skilled in the art will recognizethat the principles of the present invention may be applied to singleended or differential systems.

The RF block 102 comprises a transconductor 120, which receives theinput signal RF_(in) in the form of a differential voltage and generatesdifferential output signals that are coupled to the inputs of a mixercore 122 through a series combination of a resistor R and a capacitor C.The output of the transconductor 120 illustrated in dashed lines areinputs to the other mixer core (not shown). The resistor R and capacitorC serve as current dividers to provide current to the mixer core 122 andthe other mixer (not shown). The input currents to the mixer core 122are identified in FIG. 3 as I_(RF1) and I_(RF2), respectively. It shouldbe noted that the series RC circuit is not essential to the successfuloperation of the present invention. Rather, the RC circuit is merely oneimplementation of the splitter 18 (see FIG. 1). The present invention isnot limited by the specific implementation of the splitter 18. The mixercore 122 also receives a differential local oscillator (LO) as an inputand generates a differential baseband output signal (BB OUT).

The mixer core 122 is shown in FIG. 3 using conventional symbols forschematic diagram. The mixer core may be implemented by a transistorcircuit shown at the bottom of FIG. 3 using cross-coupled transistors ina known configuration for a differential mixer. The emitters of thevarious transistors in FIG. 3 are coupled together to form first andsecond input nodes that receive the RF signal. The input nodes arebiased by bias current sources I_(B) in a known manner. In analternative embodiment, the transconductor 120 may supply sufficientbias current thus enabling the elimination of the bias current sourcesI_(B).

The transistor arrangement of the mixer core 122 illustrated in FIG. 3comprises first and second pairs of transistors whose emitters arecoupled together to form the input nodes of the mixer core 122. Theinput nodes of the mixer core 122 are driven by the currents I_(RF1) andI_(RF2), respectively. Also illustrated at the input nodes of the mixercore 122 in FIG. 3 are voltages V_(E1) and V_(E2), respectively. Asthose skilled in the art can appreciate, the non-linear operation of thetransistors result in a second order non-linearity of the input signalwhich is present at the input nodes of the mixer core 122. Thisnon-linear component is represented by the voltage V_(E1) and V_(E2) atthe input nodes of the mixer core 122. In the embodiment illustrated inFIG. 3, there is no need for an external squaring circuit, such as thesquaring circuit 106 illustrated in FIG. 2. Rather, the system 100relies on the second order nonlinear signal inherently generated withinthe mixer core 122. The current I_(RF1) and I_(RF2) may be thought of asinputs to a squaring circuit (e.g., the squaring circuit 106 of FIG. 2)while the voltage V_(E1) and V_(E2) may be considered as outputs of thesquaring circuit. The advantage of the implementation in FIG. 3 is thatthe squaring function is an inherent byproduct of the mixer core 122 andrequires no additional circuitry (e.g., the squaring circuit 106) togenerate the squared term used by the compensation branch 104. A furtheradvantage of the implementation illustrated in FIG. 3 is that thesquared signal is generated by the mixer core 122 itself, which is alsothe source of the nonlinearity within the mixer core that results in theundesirable IM2 signal (represented in FIG. 2 as i_(IM2)). Thus, thecompensation current generated by the compensation branch 104 in FIG. 3advantageously tracks the nonlinear signal generated within the mixercore 122. Other components within the RF block 102 may be also serve asthe source of the second order nonlinear signal. For example, thetransconductor 120 may generate the second order nonlinear signal.

FIG. 3 also illustrates an implementation of the compensation branch104. Coupling resistors couple the RF currents I_(RF1) and I_(RF2) to again stage 126. The output of the gain stage 126 is coupled to avariable attenuator 128 which generates calibration currents I_(IM2cal1)and I_(IM2cal2).

The calibration current can be written as:i _(IM2cal) =I _(IM2cal1) −I _(IM2cal2) =α·g _(mcal)·ν_(E) =α·g_(mcal)·α_(2core) ·I _(RF) ²  (7)

which is of the desired form.

Using the emitter node of the mixer core 122 as the IM2 source for thecalibration is desirable because, from a simplified point of view, theIM2 generated by the mixer cores can be explained as the strong IM2signal present on the emitter node leaking unequally to the two outputsdue to mismatches in the transistors used to implement the mixer core.If no mismatch were present, the mixer core would not generate anydifferential IM2 because the emitter IM2 would leak equally to bothsides. Thus, it would be expected that the output IM2 tracks the emitterIM2 (i.e., the output IM2 would be given as a mismatch factor times theemitter IM2).

In the absence of temperature dependencies, the calibration currentcharacterized in equation (7) above would provide a suitable correctioncurrent to eliminate IM2 generated by the mixer cores. Unfortunately,simulations show that this mismatch factor is temperature dependent, andthe dependency depends on the type of mismatch (e.g., emitter resistancemismatch gives a different temperature profile than base-emittercapacitance mismatch, etc.). In practice, one type of mismatch willtypically dominate so that the temperature dependence is repeatable.Therefore, it is desirable to let the α factor have a programmabletemperature dependence. Thus, the term α in equation (7) may be alteredto include the following characteristic:

$\begin{matrix}{\alpha = {\alpha_{cal} \cdot \left( {1 + {\beta_{cal} \cdot \frac{T - T_{0}}{T_{0}}}} \right)}} & (8)\end{matrix}$

where α_(cal) and β_(cal) are programmable constants, T is temperatureand T₀ is the temperature at which calibration is performed.

The abbreviated schematic of FIG. 4 illustrates a circuit thatimplements the desired calibration function. It uses a current steeringDAC to set the calibration factor and currents I_(A) and I_(B) to setthe temperature dependence. The circuit works as follows:

Firstly, we rewrite the various currents in terms of I_(A), I_(B),I_(ref), and I_(LF):

$\begin{matrix}{{I_{DAC1} = {\frac{1}{2} \cdot \left( {1 + \alpha_{DAC}} \right) \cdot I_{ref}}}{I_{DAC2} = {\frac{1}{2} \cdot \left( {1 - \alpha_{DAC}} \right) \cdot I_{ref}}}{I_{o1A} = {\frac{1}{2} \cdot \left( {1 + \alpha_{o}} \right) \cdot I_{LF1}}}{I_{o2A} = {\frac{1}{2} \cdot \left( {1 - \alpha_{o}} \right) \cdot I_{LF1}}}{I_{o1B} = {\frac{1}{2} \cdot \left( {1 + \alpha_{o}} \right) \cdot I_{LF2}}}{I_{o2B} = {\frac{1}{2} \cdot \left( {1 - \alpha_{o}} \right) \cdot I_{LF2}}}{I_{2a} = {\frac{1}{2} \cdot \left( {1 + \alpha_{2}} \right) \cdot I_{B}}}{I_{2b} = {\frac{1}{2} \cdot \left( {1 - \alpha_{2}} \right) \cdot I_{B}}}} & (9)\end{matrix}$

Observing that I₄=0.5·(I_(B)−I_(A)), we additionally have:

$\begin{matrix}{I_{3a} = {{I_{2a} - I_{4}}\mspace{31mu} = {{{\frac{1}{2} \cdot \left( {1 + \alpha_{2}} \right) \cdot I_{B}} - {\frac{1}{2} \cdot \left( {I_{B} - I_{A}} \right)}}\mspace{31mu} = {\frac{1}{2} \cdot \left( {1 + {\frac{I_{B}}{I_{A}} \cdot \alpha_{2}}} \right) \cdot I_{A}}}}} & (10)\end{matrix}$and similarly

$\begin{matrix}{I_{3b} = {\frac{1}{2} \cdot \left( {1 - {\frac{I_{B}}{I_{A}} \cdot \alpha_{2}}} \right) \cdot I_{A}}} & (11)\end{matrix}$

Using the translinear principle, in which certain products of currentsmay be equated to other products of currents, we find that:I _(o1A) ·I _(2b) =I _(o2A) ·I _(2a) I _(o1B) ·I _(2b) =I _(o2B) ·I_(2a) I _(DAC1) ·I _(3b) =I _(DAC2) ·I _(3a)  (12)

and with the definitions provided by equations (9) and the translinearequations (12), equations (10) and (11) reduce to:

$\begin{matrix}{{{\left( {1 + \alpha_{0}} \right) \cdot \left( {1 - \alpha_{2}} \right)} = {\left( {1 - \alpha_{0}} \right) \cdot \left( {1 + \alpha_{2}} \right)}}{{\left( {1 + \alpha_{DAC}} \right) \cdot \left( {1 - {\frac{I_{B}}{I_{A}} \cdot a_{2}}} \right)} = {\left( {1 - \alpha_{DAC}} \right) \cdot \left( {1 + {\frac{I_{B}}{I_{A}} \cdot a_{2}}} \right)}}} & (13)\end{matrix}$

from which we see

$\begin{matrix}{{\alpha_{o} = \alpha_{2}}{{\frac{I_{B}}{I_{A}} \cdot \alpha_{2}} = \alpha_{DAC}}} & (14)\end{matrix}$

and thus

$\alpha_{o} = {\frac{I_{A}}{I_{B}}\alpha_{DAC}}$

Hence, the IM2 compensation current is given as

$\begin{matrix}\begin{matrix}{{I_{01} - I_{02}} = {\left( {I_{o1A} + I_{o2B}} \right) - \left( {I_{o2A} + I_{o1B}} \right)}} \\{= {\left( {I_{o2B} - I_{o1B}} \right) - \left( {I_{o2A} - I_{o1A}} \right)}} \\{= {{{{\alpha_{0} \cdot \left( {I_{LF1} - I_{LF2}} \right)}I_{o1}} - I_{o2}} = {g_{m} \cdot v_{E} \cdot \alpha_{DAC} \cdot \frac{I_{A}}{I_{B}}}}}\end{matrix} & (15)\end{matrix}$

The desired temperature variation can be implemented by letting thecurrent I_(B) be a bandgap-referenced current, and the current I_(A) bea combination of bandgap and proportional to absolute temperature(PTAT):I _(A) =I _(BG)·(1−β_(cal))+I _(PTAT)·β_(cal)I _(B)=α_(B) ·I _(BG)  (16)I _(PTAT)(T ₀)=I _(BG)

This can be done very easily using programmable current mirrors, and wethen obtain the desired function,

$\begin{matrix}{{{I_{o1} - I_{o2}} = {g_{m} \cdot \frac{\alpha_{DAC}}{\alpha_{B}} \cdot \left( {1 + {\beta_{cal} \cdot \frac{T - T_{0}}{T_{0}}}} \right) \cdot v_{E}}}\;} & (17)\end{matrix}$

It should be noted that the form of equation (17) is similar to that ofequation (8) above. Thus, FIG. 4 provides a circuit implementation ofthe compensation branch 104. It should be noted that the gain stage 126has differential inputs. One input is coupled, via two resistors, to theRF inputs of the mixer core 122 (see FIG. 3). Due to the switchingcurrents of the transistors in the mixing core 122, the signal providedto the input of the gain stage via the resistors contains both AC and DCcomponents. The signal V_(ref) is provided as a second input to the gainstage 126. The voltage V_(ref) has a value equivalent to the DCcomponent of the signal provided from the mixer core 122. Thiseffectively cancels out the DC component and allows the gain stage 126to amplify the AC signal only. The voltage V_(ref) may be generatedusing another mixer with no RF input and using the same local oscillator(LO) input. The transistors of the mixer (not shown) are matched to thetransistors of the mixer core 122 such that the DC signal produced bythe mixer core (not shown) matches the DC component generated by themixer core 122.

Due to the circuit topology, we must ensure that I_(B)>I_(A). Thecurrent I_(B) current must be set large enough to ensure this. This isdone through the α_(B) current mirror ratio described above.

As previously discussed, component mismatch in the mixer core 122 (seeFIG. 3) is a significant cause of IM2 distortion. Another cause of IM2distortion that should be considered is RF-to-LO coupling within themixer core 122. Due to mismatch in device capacitances etc. anattenuated version of the RF signal may get coupled to the LO port. Thissignal will be proportional to the incoming RF current i_(RF)(t) and maybe phase shifted by an amount φ_(leak).

On the LO port we may then have a signal component of the form,ν_(RFleakLO)(t)=γ_(leak) I(t)cos(ω_(RF) t+φ(t)+φ_(leak))  (18)where I(t) and φ(t) are a polar representation of i_(RF)(t) (i.e.,i_(RF)(t)=I(t)cos(ω_(RF)t+φ(t))).

The mixer core 122 will generate a mixing product between the RF signalon the LO port and the incoming RF current. Thus we obtain a signalcomponent at the mixer output as follows:i _(out) _(—) _(leak) =k _(mix)ν_(RFleakLO)(t)i _(RF)(t)  (19)where k_(mix) is the conversion gain of the mixer core. Expanding theabove expression yields:

$\begin{matrix}\begin{matrix}{{i_{out\_ leak}(t)} = {k_{mix}\gamma_{leak}{I(t)}{\cos\left( {{\omega_{RF}t} + {\phi(t)} +} \right.}}} \\{\left. \phi_{leak} \right){I(t)}{\cos\left( {{\omega_{RF}t} + {\phi(t)}} \right)}} \\{= {\frac{1}{2}k_{mix}\gamma_{leak}{I(t)}^{2}\left( {{\cos\left( \phi_{leak} \right)} +} \right.}} \\\left. {\cos\left( {{2\omega_{RF}t} + {2{\phi(t)}} + {2\phi_{leak}}} \right)} \right)\end{matrix} & (20)\end{matrix}$

As with the previous analysis, the high frequency component of equation(20) is easily removed with conventional filtering techniques and neednot be considered further. However, it is necessary to consider thelow-frequency part of equation (20), which may be represented asfollows:

$\begin{matrix}{{{i_{out\_ leakLF}(t)} = {a_{leak}{I(t)}^{2}}}{{{where}\mspace{14mu} a_{leak}} = {\frac{1}{2}k_{mix}\gamma_{leak}{{\cos\left( \phi_{leak} \right)}.}}}} & (21)\end{matrix}$

As is apparent, α_(leak) is a constant. Thus, the IM2 caused by RF-to-LOleakage can also be corrected by the described calibration method. It isstill advisable, however, to avoid RF-to-LO leakage. This can mosteffectively be done by ensuring low source impedance on the LO port atRF frequencies, (e.g., by using emitter-followers to drive the mixer LOport).

Since the IM2 is statistical in nature because of the variation incomponents and manufacturing processes, each wireless communicationdevice will require unique calibration current values. In oneimplementation, the compensation branch 104 is adjusted as part of afinal assembly process in a factory test. The process described aboveprovides sufficient correction for the IM2 current in the wirelesscommunication device.

In an alternative embodiment, the wireless communication device mayinclude additional circuitry to provide self-contained auto-calibration.The auto-calibration process can be automatically performed by thewireless device at regular intervals. An auto-calibration circuit isillustrated in the functional block diagram of FIG. 5. The functionalblock diagram of FIG. 5 comprises an antenna 140 and a duplexer 142.Those skilled in the art will appreciate that the duplexer 142 allows acommon antenna to be used for both transmission and reception of RFsignals. The output of the duplexer 142 is coupled to a low-noiseamplifier (LNA) 144. The output of the LNA 144 is coupled to a receiver146. Those skilled in the art will appreciate that the receiver 146generically describes all circuitry involved with the processing ofreceived signals. This includes the RF block 102 and its associatedcomponents.

The output of the receiver 146 is coupled to a mobile station modem(MSM) 148. The MSM 148 generically represents circuitry used for signalprocessing of the baseband signal. The MSM also processes baseband datafor transmission. Accordingly, the MSM 148 is also coupled to atransmitter 150. The transmitter 150 is intended to encompass allcircuitry involved in the modulation of baseband signals to theappropriate RF signals. The output of the transmitter 150 is coupled toa power amplifier (PA) 152. The PA 152 drives the antenna 140 via theduplexer 142 to transmit the RF signals. The operation of circuitcomponents, such as the MSM 148 and transmitter 150 are well known inthe art and need not be described in greater detail herein. The receiver146 is also a conventional component with the exception of the addedcircuitry of the compensation branch 104 (see FIG. 2).

Because CDMA is a full-duplex system, the transmitter 150 can be on atthe same time as the receiver 146. The present invention takes advantageof this capability by using the transmitter 150 to generate a testsignal on which to perform IM2 calibration. The simplified architectureillustrated in FIG. 5 takes advantage of the fact that IM2 distortiondoes not depend on the absolute frequencies of the signals, but only ontheir frequency separation. With the PA 152 and LNA 144 turned off, thetransmitter 150 can generate a signal that is routed straight to thereceiver 146 via semiconductor switches 156 and 158. The output signalfrom the transmitter 150 is attenuated through a resistive attenuator160.

The receiver 146 processes the received signal and the IM2 distortioncaused by the receiver results in baseband distortion product. The MSM148 can detect and minimize the distortion product by adjusting the IM2calibration. Those skilled in the art will recognize that thecalibration circuit of FIG. 5 may be used with any form of compensationcircuit. Thus, the auto-calibration circuit is not limited to thecompensation techniques described above. For example, theauto-calibration circuit of FIG. 5 could be used to compensate for thenoise figure, circuit gain, linearity, IM3 signals as well as IM2signals. In addition, the auto-calibration circuit of FIG. 5 may be usedfor different forms of IM2 compensation other than the circuit describedabove with respect to FIGS. 2-4. Thus, the present invention is notlimited by the specific form of compensation circuit.

The main signal generated by the transmitter 150 falls far into thestop-band of the baseband filter (not shown) and does therefore notcontribute any power at baseband. Consequently, the only power detectedby the MSM 148 is the IM2 distortion product and circuit noise. Thus,the MSM 148 can perform the IM2 calibration based on a simple powermeasurement.

In an alternative embodiment, illustrated in FIG. 6, an internal signalsource 162 within the receiver generates the desired test signal. In anexemplary embodiment, the signal source 162 generates a signal having atleast two frequency components that are spaced apart by a predeterminedfrequency. As previously discussed, the wireless receiver may besensitive to signals that are separated by a frequency of Δω_(J).

The signal source 162 is coupled to the input of the receiver 146 by aswitch 164. The switch 164 is activated only when the system 100 isplaced in an auto-calibration mode. The auto-calibration can beperformed at predetermined times, such as when the power is firstapplied to the wireless communication device. Alternatively, theauto-calibration can be performed periodically at predetermined timeintervals.

It is to be understood that even though various embodiments andadvantages of the present invention have been set forth in the foregoingdescription, the above disclosure is illustrative only, and changes maybe made in detail, yet remain within the broad principles of theinvention. Therefore, the present invention is to be limited only by theappended claims.

1. An automatic calibration circuit for a device having a receiver, thecircuit comprising: a signal source to generate a test signal; aselectively activated switch circuit to couple the signal source to areceiver input terminal when selectively activated to couple the testsignal to the receiver input terminal and thereby permit distortionreduction adjustments on the receiver; wherein the signal sourcecomprises an internal signal generator; and wherein the internal signalgenerator generates the test signal having multiple frequency componentshaving a predetermined spectral spacing, and the test signal is in astop-band of a filter of the receiver.
 2. An automatic calibrationcircuit for a device having a receiver and a transmitter, the circuitcomprising: a signal source comprising the transmitter to generate atest signal; a selectively activated switch circuit to couple the signalsource to a receiver input terminal when selectively activated to couplethe test signal to the receiver input terminal, wherein the test signalis in a stop-band of a filter of the receiver; and a transmitter controlto control an input signal to the transmitter, the transmitter controlbeing selectively activated during an auto-calibration process togenerate the test signal.
 3. The circuit of claim 2, further comprisingan attenuator coupled to a transmitter output terminal and therebygenerate an attenuated output signal, the attenuated output signal beingthe test signal coupled to the switch circuit.
 4. An automaticcalibration circuit for distortion reduction in a wireless communicationdevice having a receiver and a transmitter, the circuit comprising: asignal source comprising the transmitter to generate a test signal; aselectively activated switch circuit to couple the signal source to areceiver input terminal when selectively activated to couple the testsignal to the receiver input terminal, wherein the test signal is in astop-band of a filter of the receiver, and transmitter control tocontrol an input signal to the transmitter, the transmitter controlbeing selectively activated during an auto-calibration process togenerate the test signal.
 5. The circuit of claim 4, wherein the signalsource comprises an internal signal generator.
 6. An automaticcalibration circuit for distortion reduction in a device having areceiver means and a transmitter means, the circuit comprising: signalmeans comprising the transmitter means for generating a test signal;switch means for selectively coupling the test signal to a receivermeans input terminal when selectively activated, wherein the test signalis in a stop-band of a filter of the receiver means; and control means,selectively activated during an auto-calibration process, to permit thetransmitter means to generate the test signal.
 7. The circuit of claim6, wherein the signal means comprises an internal signal generator togenerate the test signal.
 8. A method for automatic calibration todistortion reduction for a wireless communication device having areceiver and a transmitter, comprising: generating a test signal usingthe transmitter, the transmitter being internal to the device;selectively coupling the test signal to a receiver input terminal whenselectively activated, wherein the test signal is in a stop-band of afilter of the receiver, and controlling an input signal to thetransmitter during an auto-calibration process to permit the transmitterto generate the test signal.
 9. The method of claim 8, whereingenerating the test signal comprises generating a test signal havingmultiple frequency components having a predetermined spectral spacing.10. An apparatus comprising: a signal source to generate a test signal;a receiver comprising: a squaring circuit for receiving a receiver inputsignal from a receiver input provided to an input of a mixer in thereceiver and generating a squared version of the receiver input signal;a gain stage for receiving the squared version of the receiver inputsignal and reproducing second order nonlinear distortion in thereceiver; and an output coupling the reproduced second order nonlineardistortion to an output of the receiver to generate a down-convertedbaseband signal characterized with reduced second order nonlineardistortion; and a selectively activated switch circuit to couple thesignal source to the receiver input when selectively activated to couplethe test signal to the receiver input.